Electronic apparatus and design method

ABSTRACT

The invention realizes a wireless communication module that is capable of transmitting the fundamental wave with low loss and reducing the double higher harmonic wave level to a desired level or lower as the whole module. The invention provides a front end module to be used for a wireless communication system such as cellular phone in which at least an output power amplifier, a matching circuit, and a low-pass filter are mounted on one insulating substrate and these circuits are connected in the above-mentioned order, wherein the relative phase of the double higher harmonic wave impedance between phases in view of the matching circuit side and the low-pass filter side from the connection point between the matching circuit and the low-pass filter is set in a range of 180 degrees ±90 degrees.

This is a continuation application of U.S. Ser. No. 10/170,375, filedJun. 14, 2002.

BACKGROUND OF THE INVENTION

This invention relates to a wireless communication system technique fortransmission of a fundamental wave of a transmission signal with a lowloss suppressing the higher harmonic wave to a level lower than apredetermined level, and more particularly relates to a technique thatis to be effectively used, for example, for a front end module of amulti-band type cellular phone.

A built-in system of a cellular phone is provided with an LSI having amicrocomputer, memory, and high-frequency circuit formodulation/demodulation of signal and electronic apparatus called as amodule on which a plurality of IC such as transistor element,capacitance element, and inductance element are mounted on an insulatingsubstrate formed of ceramic, which LSI and module are mounted on aprinted circuit, and the effort has been made to reduce the number ofelectronic apparatus for intensifying the packaging density. Forexample, a module on which a transmission power amplifier and animpedance matching circuit are mounted and a module on which atransmission/reception switching circuit and a diplexer are mounted havebeen used practically as the module used for a dual-band type cellularphone system, and the front end section mainly comprises these twomodules.

In the present patent specification, an integrated component on which aplurality of electronic apparatuses are mounted on an insulatingsubstrate such as ceramic substrate having printed wiring on the surfaceand inside thereof and in which electronic apparatuses are connected bymeans of the printed wiring and bonding wire so as to function asdesired is called as module because such integrated component can beregarded as one single electronic apparatus because of the functionthereof.

SUMMARY OF THE INVENTION

The cellular phone is required to made small-sized and light-weight. Itis important to reduce the number of electronic apparatus and intensifythe packaging density of a substrate to satisfy the above-mentionedrequirement. The inventors of the present invention tried to combine theabove-mentioned two modules, which have been mounted separately, intoone module.

In a cellular phone system, though the impedance of atransmission/reception antenna terminal is regulated to be 50 Ω, animpedance matching circuit (simply referred to as matching circuithereinafter) is interposed between a transmission power amplifier and anantenna terminal because an output impedance of the transmission poweramplifier is far lower than 50 Ω. On the other hand, it is required fora transmission path from the transmission power amplifier to the antennaterminal that the transmission loss of the fundamental wave is small andthe higher harmonic wave having frequencies that are integer multiple ofthe fundamental frequency should be reduced sufficiently.

For example, Japanese Published Unexamined Patent Application No. Hei11(1999)-234062 discloses a technique in which a resonance circuitformed by connecting an inductance element and a capacitance element inparallel is provided between an output power amplifier and a matchingcircuit so that the resonance point of the resonance circuit is broughtinto coincidence with the frequency of the higher harmonic wave tothereby attenuate the higher harmonic component as a technique forreducing the higher harmonic wave in the transmission system of acellular phone. As an another technique, a technique in which a low-passfilter, which is used to attenuate the higher harmonic wave, is providedat the rear end of a matching circuit, namely between the matchingcircuit and a transmission/reception switching circuit, has beenproposed.

It is desirable that the low-pass filter is disposed at the rear end ofthe matching circuit to reduce the higher harmonic wave withconsideration of jumping of the higher harmonic wave in the matchingcircuit. Based on the above, the inventors of the present inventiondecided to employ the system in which a low-pass filter including aresonance circuit was provided at the rear end of a matching circuit fordevelopment of the one combined module having an output power amplifier,a matching circuit, a low-pass filter, and a transmission/receptionswitching circuit.

The double higher harmonic wave is highest, the triple higher harmonicwave is second higher, and four-times higher harmonic wave is nexthigher, and the higher the order of the higher harmonic wave, the morethe intensity of the harmonic wave decreases in the case of the highorder harmonic wave arising from the fundamental wave supplied from theoutput power amplifier. Therefore, it is most important to reduce thedouble higher harmonic wave to shut off the higher harmonic wave.

For example, in the case of the dual-band cellular phone of GSM (GlobalSystem for Mobile Communication) and DCS (Digital Cellular System), itis standardized that the double higher harmonic wave level in the bandranging from 1760 to 1804 MHz is equal to or lower than −30 dBm for 900MHz fundamental wave GSM transmission and the double higher harmonicwave level in the band ranging from 1805 to 1830 MHz is equal to orlower than −36 dBm. Particularly, because the fundamental frequency ofDCS is 1.8 GHz, the condition near 1800 MHz is limited severely asdescribed hereinabove in order to prevent jumping of noise from GSMtransmission side to DCS reception side.

FIG. 3A shows a general example of a low-pass filter circuit including aresonance circuit comprising an inductance element L0 and a capacitanceelement C0 that is capable of transmitting the signal with a low loss.The capacitance elements C1 and C2 are served for transmitting thesignal to match with the impedance at a low loss. The resonance point ofthe low-pass filter that includes such resonance circuit is brought intocoincidence with the frequency of the double higher harmonic wave of thefundamental wave to thereby reduce the double higher harmonic wavesufficiently. It was found that the low-pass filter itself can be usedto reduce the double higher harmonic wave to a desired level or lower(for example, −30 dB) actually for the case in which a low-pass filterwas incorporated in a module. However, it was found that the doublehigher harmonic wave could not be reduced to the level of −30 dB orlower for the case in which an output power amplifier, a matchingcircuit, a low-pass filter, and a transmission/reception switchingcircuit were incorporated in a module as the whole.

It is the object of the present invention to provide a module used for awireless communication system comprising at least an output poweramplifier, a matching circuit, and a low-pass filter that is capable oftransmitting the fundamental wave with a low loss and capable ofreducing the double higher harmonic wave to a desired level or lower asthe whole module in the case that the low-pass filter comprising aresonance circuit is provided at the rear end of the matching circuit.

The above-mentioned and other objects and characteristics will beapparent from the detailed description of the present patentspecification and attached drawings.

The outline of representative inventions disclosed in the present patentapplication will be described herein under.

In detail, the present invention provides an electronic apparatus to beused for a wireless communication system in which at least an outputpower amplifier, a matching circuit, and a low-pass filter are mountedon one insulating substrate and connected in the above-mentioned order,wherein the phase of the double higher harmonic wave impedance in viewof the low-pass filter side from the connection point between thematching circuit and the low-pass filter is set in a range of 180degrees ±90 degrees with respect to the phase of the double higherharmonic wave impedance in view of the matching circuit from theabove-mentioned connection point.

According to the above-mentioned means, because the phase of the doublehigher harmonic wave impedance is rotated (changed) to the mismatchingposition by means of the low pass-filter, the double higher harmonicwave component of the fundamental wave can be reduced to a desired valueor lower while the transmission loss of the fundamental wave is beingsuppressed without insertion of a transmission line path having adesired length between the matching circuit and the low-pass filter.

Furthermore, a transmission/reception switching circuit and a branchingcircuit for branching a received signal including a plurality offrequency bands may be mounted on the front end module in addition tothe output power amplifier, the matching circuit, and the low-passfilter.

Furthermore, a low pass-filter comprising a resonance circuit having aninductance element and a capacitance element connected in parallel and acapacitance element to be connected between the input point and outputpoint of the resonance circuit and a constant potential point forreducing loss of the fundamental wave and impedance matching, namely alow-pass filter which do not have the input point side capacitanceelement but has only output point side capacitance element, ispreferably used as the above-mentioned low-pass filter.

The above-mentioned matching circuit preferably includes a resonancecircuit having a capacitance element and an inductance element connectedin parallel. Accordingly, the double higher harmonic wave component isattenuated in the matching circuit and is further attenuated by thelow-pass filter, whereby the level can be further reduced to be low.

Furthermore, for designing the above-mentioned module in which at leastan output power amplifier, a matching circuit, and a low-pass filter aremounted on one insulating substrate and connected in the above-mentionedorder, it is preferable to set the phase of the double higher harmonicwave impedance in view of the low-pass filter side from the connectionpoint between the matching circuit and the low-pass filter in a range of180 degrees ±X (X ranges within 90 degrees) with respect to the phase ofthe double higher harmonic wave impedance in view of the matchingcircuit from the above-mentioned connection point, and it is preferableto determine the X based on a value of the standard value added with apredetermined margin.

As the result, the module that is capable of attenuating the doublehigher harmonic wave as a whole can be obtained through a filter circuitthat is not so capable of attenuating the double higher harmonic wave isused, and designing is made easily and the manufacturing processcondition is rendered not restrictive. In other words, if a desiredattenuation is desired to be attained only by devising the filter, aserious load is loaded on a designer who is to develop the filter havingsuch excellent characteristic or the manufacturing process condition isrendered restrictive. However, in the present invention, because thedouble higher harmonic wave is further attenuated by rotating the phaseby means of the filter, the filter can be designed easily and themanufacturing process condition is rendered not restrictive.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram showing an example of a front end that issuitably used for a dual-band cellular phone capable oftransmission/reception in two systems, namely GSM and DCS.

FIG. 2 is a circuit diagram showing a detailed exemplary circuitstructure of one transmission section (for example, GSM) of the frontend module shown in FIG. 1.

FIG. 3A is a circuit diagram showing a general exemplary circuit of alow-pass filter including a resonance circuit, and FIG. 3B shows acircuit diagram showing an exemplary circuit of a low-pass filter usedfor a module to which the present invention is applied.

FIG. 4A is a graph showing the double wave attenuation characteristic ofthe low-pass filter shown in FIG. 3A, and FIG. 4B is a graph showing thedouble wave attenuation characteristic of the low-pass filter shown inFIG. 3B.

FIG. 5 is a Smith chart showing the phase characteristic of the filtershown in FIG. 3A and FIG. 3B.

FIG. 6 is a conceptual diagram showing a system that is structured tostudy the relation between the length of a cable used for connecting apower amplifier to an antenna selective switch and the double higherharmonic wave level.

FIG. 7 is a graph showing the relation between the double higherharmonic wave level and the frequency with changing the length of thecable in the system shown in FIG. 6.

FIG. 8 is a graph showing the relation between the cable length and therelative phase of the double higher harmonic wave impedance in thesystem shown in FIG. 6.

FIG. 9 is an explanatory diagram showing the phase of the double higherharmonic wave impedance ZANT corresponding to the cable length in viewof the antenna switch side from the connection point between the poweramplifier and the cable, which is shown in the form of Smith chart.

FIG. 10 is a graph showing the relation between the relative phase fromthe base point “0” of respective points (1) to (9) shown in FIG. 9 andthe double higher harmonic wave level.

FIG. 11 is an explanatory diagram showing the relation between the phaseof the double higher harmonic wave impedance ZANT corresponding to thecable length in view of the antenna switch side from the connectionpoint between the power amplifier and the cable and the phase of doublehigher harmonic wave impedance ZHPA in view of the power amplifier sidefrom the connection point between the power amplifier and the cable,which is shown in the form of Smith chart.

FIG. 12 is partial cross sectional perspective view showing a devicestructure in which the circuit shown in FIG. 1 is incorporated to form amodule.

FIG. 13 is a bottom view showing an exemplary structure of the back sideof a module of an example.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Preferred embodiments of the present invention will be described indetail hereinafter with reference to the drawings.

During the development of incorporation of an output power amplifier, amatching circuit, a low-pass filter, and a transmission/receptionswitching circuit into one module, the inventors of the presentinvention found that, though a low-pass filter itself could reduce thedouble higher harmonic wave to a desired level or lower in the case thatthe low-pass filter including a resonance circuit was used, the doublehigher harmonic wave could not be reduced to the desired level or loweras the whole module.

In the early stage, the inventors of the present invention attributedthe reason to aerial propagation jumping of the higher harmonic wavefrom the output power amplifier to the transmission/reception switchingcircuit side. Then, a transmission wire path (50 Ω, semirigid cable:referred to as cable hereinafter) L having a desired length wasinterpolated between the power amplifier HPA that included the matchingcircuit and the antenna selector switch ANTSW as shown in FIG. 6, andthe double higher harmonic wave level of the whole system shown in FIG.6 was measured with changing the length of the cable L.

The result is shown in FIG. 7. It was found from FIG. 7 that the doublehigher harmonic wave level was not proportional to the length of thecable and changed periodically though the level changed depending on thelength of the cable. Based on the above-mentioned fact, it was presumedthat the double higher harmonic wave reduction effect due tointerpolation of the cable was not dependent on the aerial propagationof the double higher harmonic wave but dependent on a certain phaserelation between the output power amplifier and thetransmission/reception switching circuit. To quantify the relationbetween the cable length and the phase and the periodicity, thevariation of the cable length was converted into the phase variation ofthe double wave impedance.

In detail, the double wave impedance ZANT was measured in view of thecable side from the connection between the matching circuit and thecable to thereby reveal the relative phase relation between the cablelength and the double wave impedance. The result is shown in FIG. 8.From FIG. 8, the correlation between the cable length and relative phaseof the double impedance is found clearly.

Based on the above-mentioned correlation, the double wave impedance ZANTand the double higher harmonic wave level were measured for the 1830 MHzfrequency signal with changing the cable length. The result is shown inFIG. 9 and FIG. 10.

FIG. 9 is a diagram formed by plotting the respective double waveimpedances ZANT on a Smith chart phase clockwise having the short point(Z=0) that is regarded as the reference point (phase=0), and FIG. 10 isa diagram for showing the relation between the relative phase from thereference point “0” of the respective points (1) to (9) shown in FIG. 9and the double higher harmonic wave level. In FIG. 9 and FIG. 10, (1)represents the relative phase caused at the connector on the startingend side of the cable, and (2) represent the relative phase caused atthe connector on the terminal end side of the cable. From FIG. 10, it isfound that the double higher harmonic wave level changes periodicallywith respect to the relative phase of the double wave impedance.

In the standard of dual-band type GSM and DCS, it is specified that thedouble wave level in the band ranging from 1805 to 1830 MHz should be−36 dBm or lower. Therefore, it is found from FIG. 10 that (2) to (4)and (6) to (9) excepting (1) and (5) satisfy requirement of thestandard. Furthermore, in the case that the condition of the double wavelevel of −40 dBm or lower is employed in consideration of a margin ofabout 10%, (2) and (7) to (9) satisfy requirement of the standard, and(1) and (3) to (6) do not satisfy requirement of the standard. In FIG.9, the phase region enclosed by a broken line A does not satisfy thecondition.

Next, it is found that the phase of the double wave impedance ZHPA inview of the matching circuit side from the connection point between thematching circuit and the cable is approximately 200 degrees. This isshown in FIG. 10 as the relative position of the characters a1 and a2.From the above-mentioned result, it is found that the double higherharmonic wave level is maximized at the point where the phase of thedouble wave impedance ZANT in view of the cable side from the connectionpoint between the matching circuit and the cable coincides with thephase of the double wave impedance ZHPA in view of the matching circuitside, and on the other hand the double higher harmonic wave level isminimized at the position where the one phase deviates from the otherphase by 180 degrees.

On the other hand, the double wave impedance ZHPA in view of thematching circuit side from the connection point between the matchingcircuit and the cable is plotted on a Smith chart together with therespective points (1) to (9) to obtain a position shown with a mark ▴ inFIG. 11. It is found that the phase included in a range from 90 degreesto 180 degrees from the phase of ZHPA satisfies the above-mentionedcondition “the double wave level of −40 dBm or lower” out of theabove-mentioned phases in the case that this point is employed as thereference.

Herein, based on the fact that the double wave level is optimal at theposition where the phase deviates 180 degrees from the phase of ZHPA, itis estimated that the phase range from 180 degrees to 270 degrees alsosatisfies the condition “the double wave level of −40 dBm or lower”though it is not plotted in FIG. 11. In other words, the length of thecable is designed so that the double wave impedance ZANT is included inthe range of 180 degrees ±90 degrees from the phase of ZHPA to therebysatisfy the condition.

The range of 180 degrees ±90 degrees is the condition to reduce thedouble wave level to −40 dBm or lower, and the range changes dependingon the upper limit. For example, in the case that the level is set to−45 dBm or lower to have a margin of 25%, it is found that the range isnarrowed to 180 degrees ±45 degrees in this case because only the points(8) and (9) satisfy the condition in FIG. 10.

However, when a test was carried out by use of a ceramic substratehaving a relative dielectric constant approximately similar to (9)provided with the cable to satisfy the above-mentioned phase relation,it was found that the cable length of ten and several mm was required tosatisfy the above-mentioned phase relation and the so long cable causeda trouble to miniaturize the module when an output power amplifier, amatching circuit, a low-pass filter, and a transmission/receptionswitching circuit were incorporated together to form one module. Toavoid the trouble, it was tried to adjust the phase of a low-passfilter, and such low-pass filter type was studied. As the result, it wasfound that a module that was capable of transmitting the fundamentalwave with a low loss with reduced double higher harmonic wave to a levelequal to or lower than the standard level could be realized by use of alow-pass filter as shown in FIG. 3B. As the result, the presentinvention has been accomplished.

Though it is possible to rotate the phase of the double wave impedanceby changing the constant of an inductance and capacitance with using thelow-pass filter of the circuit type shown in FIG. 3A, this method is notpreferable because the attenuation of the double wave arising from thefilter itself becomes poor.

FIG. 1 shows one exemplary front end suitably used for a dual-band typecellular phone that is capable of two types of transmission/reception,namely GSP and DCS.

In FIG. 1, ANT denotes a transmission/reception antenna fortransmitting/receiving signal wireless wave, 100 denotes a front endmodule to which the present invention is applied, 200 denotes a highfrequency processing circuit for down-converting and demodulating thefrequency of a received signal to an intermediate frequency to therebygenerate a base band signal and for modulating a received signal, 300denotes a base band circuit for converting an audio signal to a baseband signal and for converting a received signal to an audio signal,FLT1 and FLT2 denote filters for removing noise from a received signal,and LNA1 and LNA2 denote low-noise amplifiers for amplifying a receivedsignal.

For example, the filter FLT1 and the amplifier LNA1 are served for a GSMcircuit, and the filter FLT2 and LNA2 are served for aDCS circuit. Thehigh frequency processing circuit 200 comprises one or moresemiconductor integrated circuits. The base band circuit 300 comprises aplurality of LSI and IC such as DSP (Digital Signal Processor),microprocessor, and semiconductor memory.

The front end module 100 comprises output power amplifier 111 and 112for amplifying a signal received from the high frequency processingcircuit 200, impedance matching circuits 121 and 122, low-pass filters131 and 132 for attenuating the higher harmonic wave,transmission/reception switching circuits 141 and 142, capacitances 151and 152 for cutting the DC component from a received signal, a diplexer160 for branching a signal into a GSM type signal of 900 MHz band and aDCS type signal of 1.8 GHz band. These circuits and elements are mountedon one ceramic substrate to form a module.

The module used in the present example is provided with a terminal 170that is designed to have an impedance of 50 Ω to which thetransmission/reception antenna ANT is connected. The impedance of theconnection point between the impedance matching circuits 121 and 122 andthe low-pass filters 131 and 132 for attenuating the higher harmonicwave and the impedance of the connection point between the low-passfilters 131 and 132 and the transmission/reception switching circuits141 and 142 are also designed to have an impedance of 50 Ω respectively.

Furthermore, though it should be limited not necessarily, in the presentexample, the transmission/reception switching circuits 141 and 142 areswitched in response to switching control signals CNT1 and CNT2 suppliedfrom the base band circuit 300.

FIG. 2 shows a detailed exemplary circuit of one transmission section(for example, GSM) of the front end module 100 shown in FIG. 1.

In FIG. 2, Tr1 is the final end transistor of the output power amplifier111, a transmission signal is supplied to the gate terminal of thetransistor Tr1, a power voltage Vd is applied on the drain terminalthrough λ/4 transmission line path TL1 having an electric length of ¼wavelength of the fundamental wave, and the impedance matching circuit121 is connected to the connection node between the λ/4 transmissionline path TL1 and the drain terminal of the transistor Tr1. In theabove, the TL1 is not necessarily a ¼ line path but may be a coilinductance.

Though it is limited not necessarily, in the present example, theimpedance matching circuit 121 is provided with a parallel resonancecircuit PR having an inductance element and a capacitance element, amatching means IM comprising a transmission line path and a capacitanceelement, and a capacitance element DC for cutting DC noise that comesfrom the low-pass filter 131 side to the power amplifier side. Theconstant of the circuit is set so that the impedance of the output nodeof the impedance matching circuit 121 is 50 Ω. MOSFET is used as theoutput transistor Tr1 in FIG. 1, but the output transistor is notlimited to MOSFET, and a bipolar transistor, GaAsMESFET,hetero-connection bipolar transistor (HBT), or HEMT(high-electron-mobility transistor) may be used instead.

The impedance of the input terminal and output terminal of thetransmission/reception switching circuit 141 is 50 Ω, though it isdesirable that a switching circuit that does not change the phase of theimpedance in the view from the low-pass filter 131 side is used, adesigner can design a switching circuit having such characteristicgenerally, and such characteristic can be realized by use of a switchingcircuit that is used for a conventional antenna switching circuit moduleof an cellular phone. Therefore the detailed description will beomitted.

The diplexer 160 comprises the low-pass filter LPF having a parallelresonance circuit, a capacitance element connected between the nodelocated opposite to the antenna terminal 170 of the parallel resonancecircuit and the earth potential point, and a DCS cutoff capacitanceelement C11 disposed between the resonance circuit and thetransmission/reception switching circuit 141 and the high-pass filterHPF having a parallel resonance circuit, an inductance element connectedbetween the node located opposite to the antenna terminal 170 of theparallel resonance circuit and the earth potential point, and a DCcutoff capacitance element C12.

The low-pass filter 131 comprises a parallel resonance circuit having aninductance element L0 and a capacitance element C0 and a capacitanceelement C2 connected between the output point of the resonance circuitand the earth potential point for reduction of loss of the fundamentalwave and impedance matching. By using the low-pass filter of this type,the phase of the double wave impedance is rotated to a desired positionto thereby reduced the double higher harmonic wave level to −40 dBm orlower. The characteristic of this filter will be described in detailherein under.

A usual low-pass filter comprising a resonance circuit formed byconnecting an inductance element L0 and a capacitance element C0 inparallel, a capacitance element C1 connected between the input point ofthe resonance circuit and the earth potential point, and a capacitanceelement C2 connected between the output point of the resonance circuitand the earth potential point as shown in FIG. 3A. Both the capacitanceelements are served for reduction of loss of the fundamental wave andimpedance matching. The double wave attenuation characteristic of suchlow-pass filter is shown in FIG. 4A. Furthermore, the low-pass filtershown in FIG. 3B has no capacitance C1 to be located on the inputterminal side, which is provided in the case of the circuit shown inFIG. 3A. The double wave attenuation characteristic of the low-passfilter of FIG. 3B is shown in FIG. 4B. It is apparent from comparisonbetween FIG. 4A and FIG. 4B that both the filters themselves canattenuate the double higher harmonic wave approximately in the sameextent.

On the other hand, FIG. 5 shows the phase characteristic of thesefilters shown in FIG. 3A and FIG. 3B in the form of Smith chart. In FIG.5, LPFa shows the phase characteristic of the filter shown in FIG. 3A,and LPFb shows the phase characteristic of the filter shown in FIG. 3Brespectively. A mark ▾ shows the phase position of the double wave (1.8GHz) impedance for the filter shown in FIG. 3A, and a mark ∇ shows thephase position of the double wave impedance for the filter shown in FIG.3B.

From FIG. 5, it is apparent that the filter shown in FIG. 3B can rotatethe double wave impedance phase of the filter shown in FIG. 3A byapproximately 90 degrees more. Thereby, the filter circuit is designedso as not to phase-match with the double higher harmonic waveintentionally, and the double higher harmonic wave can be reduced duringpassage through the filter.

As described in SUMMARY OF THE INVENTION, the module in which the usuallow-pass filter as shown in FIG. 3A connected at the rear end of thematching circuit is used is insufficient to reduce the double higherharmonic wave level to −30 dBm or lower, but the module described in theexample shown in FIG. 1 in which the filter shown in FIG. 3A is used canrotate the double wave impedance phase by approximately 90 degrees moreto thereby reduce the double higher harmonic wave level to −30 dBm orlower.

The impedance phase of the fundamental wave (900 MHz) of the filtershown in FIG. 3A is positioned at the center (the point indicates noloss with 50 Ω) as shown with the mark ▴ in Smith chart shown in FIG. 5.On the other hand, the impedance phase of the fundamental wave of thefilter shown in FIG. 3B is positioned at the mark Δ shown in Smith chartshown in FIG. 5. Because both the filters shown in FIG. 3A and in FIG.3B cause scarce insertion loss of the fundamental wave inherently, theattenuation action due to phase mismatching of the impedance isnegligibly small.

Because the filter shown in FIG. 3B has no capacitance C1 though thefilter shown in FIG. 3A has both capacitances C1 and C2 served for lossreduction, the transmission loss in the case of the filter shown in FIG.3B is slightly larger than that in the case of the low-pass filter shownin FIG. 3A as it is apparent in comparison between FIG. 4A and FIG. 4B.However, because the increment is relatively small, the totalperformance of the module of the present example is regarded to bebetter than that of the comparative module in consideration of theadvantage that the double higher harmonic wave level is reduced to −30dBm or lower despite the reduction of transmission loss. Accordingly,the filter of circuit type shown in FIG. 3B is employed.

In the case of the conventional front end comprising two modules, thedouble higher harmonic wave is reduced by changing the layout of the twomodules to change the length of the transmission line path (wiring) forconnecting between the modules. However, in this case, because thelayout of the modules is designed relying upon intuition of a designerand there is no effective means for designing the layout of the modulescorrectly, the length of the transmission line path is too longinevitably and the long transmission line path prevents the system frombeing miniaturized.

On the other hand, the module can be miniaturized by applying thepresent example easily, and the double higher harmonic wave is reducedreliably. The change of the layout of the two modules of theconventional front end comprising the two modules means the change ofthe length of the transmission line path (wiring) for connecting betweenthe modules, and the change of the length of the transmission line pathresults in the rotation of the impedance phase as in the case of thefilter circuit of the present example. However, the conventional designdoes not involve the active intention to rotate the impedance phase bychanging the transmission line path (wiring) The reason is that thelength of the transmission line can be designed not based on experiencebut based on calculation if the transmission line path is changed withactive intention.

FIG. 12 shows a device structure in which the circuit shown in FIG. 1 isincorporated in the form of a module. FIG. 12 is a diagram that does notshows the structure of the front end module of the present examplecorrectly, but shows the structural diagram from which the detail isomitted to show the outline.

As shown in FIG. 12, the body 10 of the module of the present examplecomprises a plurality of dielectric plates 11 such as ceramic plateconsisting of alumina combined into one piece. On the front surface andback surface of each dielectric plate 11, a conductive layer 12consisting of conductive material such as copper plated with gold onwhich a desired pattern is formed is provided. 12 a denotes a wiringpattern comprising a conductive layer 12. Furthermore, a hole 12 calledas a though hole is formed on each dielectric plate 11 to connectbetween conductive layer 12 or wiring pattern together on the front andback surfaces of each dielectric plate 11, and conductive material isfilled in the hole.

In the case of the exemplary module shown in FIG. 12, six dielectricplates 11 are laminated, conductive layers 12 are formed on the almostentire surface of the back side of the first layer, third layer, andsixth layer from the top, which are served as the ground layer forsupplying the earth potential GND respectively. Conductive layers 12provided on the front and back surfaces of other dielectric plates 11are served for the transmission line path. The width of the conductivelayers 12 and the thickness of the dielectric plates 11 are designed sothat the impedance of the transmission line path is adjusted to be 50 Ω.

A rectangular hole is formed on the first to third dielectric plates 11to dispose GSP system power amplifier IC21 and DCS system poweramplifier IC22. Each IC is inserted into the inside of the hole andfixed on the bottom of the hole with binder 14. Holes 15 called as viahole are formed on the fourth dielectric plate 11 located at theposition corresponding to the bottom of the hole and on dielectricplates 11 laminated under the fourth dielectric plate 11, and conductivematerial is filled in the holes. The conductive material filled in thevia holes is served to transfer the heat generated from the IC21 andIC22 to the lowermost conductive layer to dissipate the heat and improvethe thermal efficiency.

Electrodes on the top surface of the IC21 and IC22 and the predeterminedwiring pattern 12 a are connected electrically by means of bonding wire31. Furthermore, on the surface of the first layer dielectric plate 11,a plurality of chip-type electronic apparatuses 32 such as capacitances,resistive elements, inductances are mounted to form the above-mentionedmatching circuit 121 and filter circuit 131. Otherwise, these elementsmay be formed in the internal of the substrate by use of conductivelayers on the front and back surfaces of dielectric plates 11 instead ofuse of the electronic apparatus.

The module has an external terminal served for mounting the module ofthe present example on a printed wiring board by connecting electricallyeach other. The external terminal is an electrode pad 41 comprising aconductive layer that is formed in a predetermined shape, and theexternal terminal is disposed on the back surface of the module body 10as shown in FIG. 13. The external terminal is structured so as to bemounted on the printed wiring board with interposition of a solder ballbetween the electrode pad and the corresponding portion located on theprinted wiring board of the system (a portion of the wiring orconductive layer connected to the wiring).

The layout and the configuration of the electrode pad 41 shown in FIG.13 only shows an example, and that is by no means limited to theexample. Furthermore, the conductive layer 12 that is served as theground layer for supplying the earth potential is formed on almostentire region excepting the surface of the electrode pad 41 as describedhereinabove in FIG. 13.

The invention accomplished by the inventors of the present invention hasbeen described based on the example in detail, however, the presentinvention is by no means limited to the above-mentioned example, and asa matter of course various modifications may be applied withoutdeparting from the sprit and the scope of the present invention.

For example, in the case of the module shown in FIG. 1, (1) output poweramplifiers 111 and 112, (2) impedance matching circuits 121 and 122, (3)low-pass filters 131 and 132, (4) transmission/reception switchingcircuits 141 and 142, (5) DC cut-off capacitances 151 and 152, and (6)diplexer 160 are mounted on one ceramic substrate to fabricate themodule. However, the present invention may be applied to the case inwhich (1) output power amplifiers 111 and 112, (2) impedance matchingcircuits 121 and 122, and (3) low-pass filters 131 and 132 are used tofabricate one module, or the case in which (1) output power amplifiers111 and 112, (2) impedance matching circuits 121 and 122, (3) low-passfilters 131 and 132, and (4) transmission/reception switching circuits141 and 142 are used to fabricate one module. In all the cases, the sameeffect is obtained.

Furthermore, some conventional high frequency power amplifier modulesare provided with a coupler for detecting the output level of the poweramplifier and an APC (Automatic Power Control) circuit for controllingthe bias voltage of the output transistor element based on the outputsupplied from the coupler on the rear end of the power amplifier so thatthe sufficient output power for communication can be obtained. Also inthe case of the front end module of the present example, similar couplerand APC may be incorporated to form one module.

Furthermore, in the example shown in FIG. 2, only one low-pass filter131 is connected to the rear end of the matching circuit 121, butadditional one low-pass filter may be connected to the rear end of theabove-mentioned low-pass filter to attenuate the higher harmonic wave ofthird order or higher. Furthermore, the resonance circuit PR is providedin the matching circuit 121 in the example shown in FIG. 2, but amatching circuit comprising a transmission line and capacitance elementfrom which the resonance circuit is omitted may be used. Atransmission/reception switching circuit 141 having the structure otherthan that shown in FIG. 2 may be used.

The case in which the present invention is applied to the front endmodule that is effectively used for the dual-band cellular phone that iscapable of transmission/reception in two systems, namely GSM and DCS,which is the application field of the background for inventing thepresent invention accomplished by the inventors, is describedhereinbefore. However, application of the present invention is by nomeans limited to the case, and the present invention may be applied tovarious wireless communication systems such as multi-band cellularphones and mobile telephones that are capable of transmission/receptionin three or more systems.

The effect obtained by applying the typical invention out of inventionsdisclosed in the present patent application is described herein under.

The invention provides a wireless communication module that is capableof transmitting the fundamental wave with low loss and reducing thedouble higher harmonic wave level to the desired level or lower as thewhole module in the case that a low-pass filter comprising a resonancecircuit is provided on the rear end of a matching circuit. Furthermore,one module is sufficient for use thereby through two modules are usedconventionally, and a miniaturized wireless communication system such ascellular phone is realized by using this type of module.

1. An electronic apparatus to be used for a wireless communicationsystem in which at least an output power amplifier, a matching circuit,and a low-pass filter are mounted on one insulating substrate andconnected in this order, wherein the phase of the double higher harmonicwave impedance in view of the low-pass filter side from the connectionpoint between the matching circuit and the low-pass filter is set in arange of 180 degrees ±90 degrees with respect to the phase of the doublehigher harmonic wave impedance in view of the matching circuit from theconnection point.
 2. The electronic apparatus according to claim 1,wherein the low-pass filter comprises a resonance circuit having acapacitance element and an inductance element connected in parallel anda capacitance element connected between the output point of theresonance circuit and a constant potential point.
 3. The electronicapparatus according to claim 2, wherein the matching circuit isstructured including a resonance circuit comprising a capacitanceelement and inductance element connected in parallel.
 4. The electronicapparatus according to claim 3, wherein a transmission/receptionswitching circuit for switching the signal between a transmission signaland a received signal is additionally mounted on the insulatingsubstrate.
 5. The electronic apparatus according to claim 1, wherein aset of the output power amplifier, a matching circuit, and a low-passfilter is provided correspondingly to each frequency band of two or morefrequency bands, and the double higher harmonic wave of any one of thefrequency bands is in any one of other frequency bands.
 6. Theelectronic apparatus according to claim 5, wherein a branching circuitfor branching a received signal into the two or more frequency bands isadditionally mounted on the insulating substrate.
 7. An electronicapparatus used for a wireless communication system that communicates byuse of two or more frequency bands, wherein at least a set of an outputpower amplifier, a matching circuit, and a low-pass filter is providedcorrespondingly to each frequency band, the sets are mounted on oneinsulating substrate, and the output power amplifier, the matchingcircuit, and the low-pass filter of each set are connected in thisorder, and wherein the phase of the double higher harmonic waveimpedance in view of the low-pass filter side from the connection pointbetween the matching circuit and the low-pass filter is deviated fromthe phase matching position by means of the low-pass filter.
 8. Theelectronic apparatus according to claim 7, wherein the low-pass filtercomprises a resonance circuit having a capacitance element and aninductance element connected in parallel and a capacitance elementconnected between the output point of the resonance circuit and aconstant potential point.
 9. The electronic apparatus according to claim8, wherein the matching circuit is structured including a resonancecircuit comprising a capacitance element and inductance elementconnected in parallel.
 10. The electronic apparatus according to claim9, wherein a transmission/reception switching circuit for switching thesignal between a transmission signal and a received signal isadditionally mounted on the insulating substrate.
 11. The electronicapparatus according to claim 10, wherein a branching circuit forbranching a received signal into a plurality of frequency bands isadditionally mounted on the insulating substrate. 12-15. (canceled) 16.A design method for designing an electronic apparatus in which at leastan output power amplifier, a matching circuit, and a low-pass filter aremounted on one insulating substrate and connected in this order, whereinthe phase of the double higher harmonic wave impedance in view of thelow-pass filter side from the connection point between the matchingcircuit and the low-pass filter is set in a range of 180 degrees ±X (Xranges within 90 degrees) with respect to the phase of the double higherharmonic wave impedance in view of the matching circuit from theconnection point, and wherein the X is determined based on the standardvalue to which a predetermined margin is added.
 17. The design methodfor designing an electronic apparatus according to claim 16, wherein theX is 90 degrees in the case that the margin is approximately 10% of thestandard value.
 18. The design method for designing an electronicapparatus according to claim 16, wherein the X is 45 degrees in the casethat the margin is approximately 25% of the standard value.